The present invention relates to a technique for facilitating reduction in noise and a design of a loop filter relative to a transmitter which has a high frequency power amplifier and performs a control of an output power by a feedback control, and particularly to a technique effectively applied to a transmitter having a phase control loop and a technique effectively applied to an amplitude control loop for performing a phase modulation and an amplitude modulation and a wireless communication apparatus using this transmitter such as a mobile phone.
In a mobile phone, needs of a high-speed data communication in addition to a speech communication have been increased in recent years. In GSM (Global System for Mobile communications) that is one of European Mobile Communications Standards, on the basis of a conventional system using a GMSK (Gaussian Minimum Shift Keying) modulation architecture, GPRS (General Packet Radio Service) for speeding up a data communication has been decided by allowing a plurality of time slot transmissions for one time slot transmission in the GSM with multiplexing TDMA (Time Division Multiple Access) architecture. Further, in order to realize a data speed exceeding the GPRS, a standardization of EDGE (Enhanced Data for GSM Evolution) using 8-PSK as a modulation architecture has been made.
Since an amplitude of a GMSK modulated signal is constant, as a transmission architecture of a mobile phone for GSM, an offset PLL architecture for outputting a signal having a constant amplitude is generally used. Further, in the offset PLL architecture, since an input amplitude is constant, as a power amplifier for amplifying a signal with a predetermined gain, a high efficient nonlinear power amplifier is widely used. The operating principle of the transmitter of the offset PLL architecture is described in, for example, IEEE journal of solid-state circuits, Vol. 32, No. 12, December 1997, “A 2. 7-V GSM RF transceiver IC”, pp. 2089-2096.
On the other hand, since the amplitude of the modulated signal in the 8-PSK modulation of the EDGE system is not constant, a linear transmission capable of being transmitted without distortion of not only an input signal phase but also an amplitude is required for the transmission architecture. As a way to realize the linear transmission, two architectures have been known. The first architecture is a mixer architecture for performing a frequency conversion by a mixer, where a linear power amplifier is employed. Details of the mixer architecture are described in, for example, “RF MICROELECTRONICS”, pp. 149-155 by Behard Razavi, PRENTICE HALL PTR Press. The second architecture is an architecture in which a nonlinear power amplifier is employed and a distortion compensation is applied thereto, and thereby a high efficient nonlinear power amplifier can be used. As examples of the above-mentioned architecture, there are a polar-loop architecture, a Cartesian-loop architecture, a Predistortion architecture, and the like.
In the EDGE system, nine modes of MSC1 to MSC9 are, however, prescribed according to an amount of transmission data, and the modes each have a different error correcting code architecture, and a mobile phone for EDGE has to be configured so as to be capable of operating in all the modes. The modes of MSC1 to MSC4 among the nine modes of MSC1 to MSC9 relate to the GMSK modulation while the modes of MSC5 to MSC9 relate to the 8-PSK modulation. In other words, the mobile phone for EDGE has to be a mobile phone for dual mode capable of performing two modulations of the GMSK modulation and the 8-PSK modulation.
In order to realize the dual mode transmission, when the first architecture using the above-mentioned mixer is applied to both the GMSK modulation and the 8-PSK modulation, there is an advantage of reduction in area by sharing a circuit while there is a disadvantage of reduction in power efficiency because of use of the linear power amplifier. On the other hand, when the above-mentioned offset PLL architecture is employed for the GMSK modulation and the first architecture is employed for the 8-PSK modulation, there is an advantage of the case where high power efficiency is obtained while there is a disadvantage of the case where an area increases because a circuit such as a power amplifier or the like cannot be used in common therewith.
Therefore, the second architecture capable of using the nonlinear power amplifier is preferable in that the power efficiency is improved while the polar-loop architecture in the second architecture is particularly advantageous in that many circuits can be used in common with the offset PLL architecture.
FIG. 7 is a diagram showing a conventional example of a transmitter for polar-loop architecture. The above-mentioned transmitter has a phase loop and an amplitude loop.
The amplitude loop is configured with a nonlinear power amplifier 200 having an I/O terminal and an amplitude control terminal; signal branching means 201 for branching an output signal of the nonlinear power amplifier 200 into a first and second outputs; an attenuator 102 for attenuating the signal branched by the signal branching means 201; a mixer 103 to which the signal attenuated in the attenuator 102 is supplied; a voltage-controlled oscillator (VCO) 104 generating a local signal for causing the mixer 103 to perform a frequency conversion operation; a filter 105 for suppressing undesired harmonics included in the output of the mixer 103; an amplitude detector 130 for detecting an amplitude difference between a feedback signal FB and a reference modulated signal (MOD); and a low pass filter 111.
A signal (OUT) being output from the nonlinear power amplifier 200 of the above-mentioned transmitter is output to an antenna (not shown) via the signal branching means 201. An output signal of the low pass filter 111 is supplied to the amplitude control terminal of the nonlinear power amplifier 200 as an output control signal VAPC, and an output amplitude of the nonlinear power amplifier 200 is controlled such that the amplitudes of the feedback signal FB and the reference modulated signal (MOD) become equal. Further, an output signal (carrier) φTX of the transmission oscillator (VCO) 114 is input into the input terminal of the nonlinear power amplifier 200. The amplitude of the signal (carrier) φTX supplied to the input terminal of the nonlinear power amplifier 200 is constant.
FIG. 8 shows characteristics of the output amplitude to the output control signal VAPC of the nonlinear power amplifier 200 shown in the polar-loop architecture in FIG. 7. A linear area shown in FIG. 8 is used as an operating area of the nonlinear power amplifier 200. Since the filter 105 is directed for suppressing the undesired harmonics included in the output of the mixer 103 and is designed such that a band thereof is generally wider than a loop band of the above-mentioned amplitude loop, the band and a phase margin of the amplitude loop is determined according to the characteristics of the low pass filter 111.
On the other hand, the phase loop in the polar-loop architecture shown in FIG. 7 is configured with the nonlinear power amplifier 200; the signal branching means 201; the attenuator 102; the mixer 103; the local VCO 104; the filter 105; a phase detector 140 for detecting a phase difference between the feedback signal FB and the reference modulated signal (MOD); the low pass filter 113; and the transmission oscillator (VCO) 114, wherein the oscillation operation of the transmission oscillator (VCO) 114 is controlled such that the phases of the feedback signal FB and the reference modulated signal (MOD) are coincided with each other.
As described above, separate control loops are provided for an amplitude component and a phase component of the reference modulated signal (MOD) so that while a modulation spectrum of the reference modulated signal MOD is stored in the output OUT, a center frequency thereof is converted into a desired frequency. The control of the desired frequency described above is performed by setting a frequency of the local VCO 104. Note that details of the operating principle of the polar-loop architecture is described in, for example, “HIGH-LINEARITY RF AMPLIFIER DESIGN”, pp. 161-164 by PETER B. KENINGTON, Artech House Press.
In the amplitude loop of the transmitter for polar-loop architecture, in order to compensate for a distortion component generated in the nonlinear power amplifier 200, it is, however, necessary that an open loop transfer function Ho of the above-mentioned amplitude loop has a sufficiently large gain in a low frequency area. Therefore, the design is generally made such that a transfer function F of the low pass filter 111 has a pole (DC pole) at 0 Hz. General design formulas of the transfer function F in the case of one pole (Type I) and of the transfer function F in the case of two poles (Type II) are expressed in formula 1 and formula 2, respectively. In the formulas, each of A and B is constant.
                                 F          =                      A            S                                                (                      Formula            ⁢                                                  ⁢            1                    )                                              F          =                                    B              ·                              (                                  s                  +                                      w                    z                                                  )                                                                    s                2                            ·                              (                                  s                  +                                      w                    p                                                  )                                                                          (                      Formula            ⁢                                                  ⁢            2                    )                    
Formula 1 is a transfer function having one perfect integrator, in which the amplitude loop having the transfer function is stable because a phase thereof is not shifted 90 degrees or more. On the other hand, the filter expressed with formula 2 is configured with one perfect integrator and a secondary passive filter having lag-lead characteristics, in which the gain in the low frequency is increased and the phase margin is increased. The frequency characteristic of each open loop transfer function Ho of Type I and Type II at a loop band of 1.8 MHz is shown in FIG. 9. The zero point and the poles in Type II are arranged symmetrically with respect to respective loop bands, and the phase margin is designed to be 68 degrees. Generally, the phase margin is designed to be 45 degrees or more.
As can be seen from FIG. 9, since the gain in the low frequency area is larger in Type II than in Type I, Type II is more advantageous than Type I in that distortion is reduced and the modulated precision is improved. Further, the gain in a frequency having a higher loop band than the loop band (of 1.8 MHz or more) is lower in Type II than in Type I, which means that the suppression degree of noise generated in the above-mentioned amplitude loop is large, and so Type II is more advantageous for a mobile phone which requires low noise characteristics. Therefore, in the transmitter for polar-loop architecture, it is more advantageous that the low pass filter 111 on the amplitude loop is designed with Type II due to the distortion compensation and the noise suppression.